The usage of light-emitting diodes (LEDs) to provide illumination is increasing rapidly as the cost of LEDs decrease and the endurance of the LEDs increases to cause the overall effective cost of operating LED lighting products to be lower than incandescent lamps and fluorescent lamps providing equivalent illumination. Also, LEDs can be dimmed by controlling the current through the LEDs because LEDs are current driven devices. The current through a plurality of LEDs in a lighting device must be controlled tightly in order to control the illumination provided by the LEDs. Although direct sensing of the current in LEDs is desirable, certain safety requirements (e.g., UL safety standards) require the secondary of an LED lighting device to be electrically isolated from the primary (line and neutral side) of the lighting device. The isolation requirement causes conventional sensing techniques in the secondary side to be costly because of the extra components required to produce signal conversion and to provide isolation.
The foregoing problem is illustrated by a typical LED driver circuit 100 in FIG. 1. The driver circuit includes a DC-to-AC inverter 102 on a primary side of the circuit and an AC-to-DC converter 104 on a secondary side of the circuit.
The DC-to-AC inverter 102 receives a voltage from a DC source 110, which may be a DC voltage supply that produces a DC voltage from an AC source (not shown). In the illustrated driver circuit, the DC source is illustrated as a conventional battery. The voltage from the DC source is provided on a VRAIL supply line 112. The voltage on the VRAIL supply line is referenced to an input (primary side) ground reference 114. A first semiconductor switch (e.g., a power metal oxide semiconductor field effect transistor (MOSFET) or a bipolar junction transistor (BJT) 120 has a first terminal connected to the VRAIL supply line and has a second terminal connected to a common switching node 122. A second semiconductor switch (MOSFET or BJT) 124 has a first terminal connected to the common switching node and has a second terminal connected to the input ground reference. Together, the two switches operate as a half-bridge circuit 126 to produce a switched DC voltage on the common switching node.
The control terminal (e.g., gate of a MOSFET or base of a BJT) of the first switch 120 is connected to a first output (GH) 132 of an integrated circuit switch controller 130. The control terminal of the second switch 124 is connected to a second output (GL) 134 of the switch controller. In one embodiment, the switch controller comprises a UBA2014 driver integrated circuit (IC) commercially available from NXP B.V. of Eindhoven, Netherlands. Other switch controllers may be used. The switch controller operates in a conventional manner to turn on the first switch to couple the common switching node 122 to the VRAIL supply line 112. The switch controller then turns on the second switch to couple the common switching node to the input ground reference 114. When one of the switches is turned on, the other switch is turned off. The two switches are turned on and off at a selected repetition rate (operating frequency) and with selected duty cycles to produce a voltage on the common switching node that alternates between the VRAIL voltage and ground.
In the illustrated embodiment, the switch controller 130 varies the operating frequency in response to a control signal on a control (CTL) input of the switch controller. For example, the magnitude of the control signal may be varied in a first magnitude direction to increase the operating frequency and may be varied in a second magnitude direction to decrease the operating frequency. In the embodiment of FIG. 1 using the UBA2014 driver IC as the switch controller, the control (CTL) input is the CSW input of the IC, which is an input to an internal voltage controlled oscillator. The switch controller is responsive to an increased voltage on the control input to decrease the switching frequency and is responsive to a decreased voltage on the control input to increase the switching frequency. The source of the control signal is discussed below.
The common switching node 122 of the half-bridge circuit 126 is connected to a resonant tank circuit 140 including a resonant circuit inductor 142 and a resonant circuit capacitor 144. A first terminal of the resonant circuit inductor is connected to the common switching node of the half-bridge circuit. A second terminal of the resonant circuit inductor is connected a first terminal of the resonant circuit capacitor at a resonant tank node 146. The second terminal of the resonant circuit capacitor is connected to the input ground reference 114.
The resonant tank circuit 140 further includes a DC-blocking capacitor 150 and the primary winding 162 of an isolation transformer 160. One terminal of the DC-blocking capacitor is connected to the second terminal of the resonant circuit inductor 142 and to the first terminal of the resonant circuit capacitor 144 at the resonant tank node 146. A second terminal of the DC-blocking capacitor is connected to a first terminal 164 of the primary winding of the isolation transformer. A second terminal 166 of the primary winding of the isolation transformer is connected to the input ground reference 114. Thus, the DC-blocking capacitor is connected in series with the primary winding of the isolation transformer.
As described above, the components on the primary side of the isolation transformer 160 operate as the DC-to-AC inverter 102 to convert the DC power provided by the DC source 110 to an AC voltage applied to the primary winding 162 of the isolation transformer.
In the illustrated embodiment, the isolation transformer 160 includes a secondary winding 170 having a first terminal 172 and a second terminal 174. The first terminal of the secondary winding is connected to the anode of a first fast rectifier diode 180 and to the cathode of a second fast rectifier diode 182. The second terminal of the secondary winding is connected to the anode of a third fast rectifier diode 184 and to the cathode of a fourth fast rectifier diode 186. The cathodes of the first and third fast rectifier diodes are connected together at a positive (+) output node 190. The anodes of the second and fourth fast rectifier diodes are connected together at a negative (−) output node 192. The negative output node is connected to an output (secondary side) ground reference 194. The output ground reference is electrically isolated from the input ground reference 114 by the isolation transformer. Thus, the output ground reference may float with respect to the input ground reference as required by UL safety standards.
The four rectifier diodes 180, 182, 184, 186 form a full-wave bridge rectifier 198. The full-wave bridge rectifier operates in a conventional manner to rectify the AC voltage from the secondary winding 170 of the isolation transformer 160 to generate a rectified DC voltage between the positive output terminal 190 and the negative output terminal 192.
A first output filter capacitor 200 has a first terminal connected to the positive output terminal 190 and has a second terminal connected to the negative output terminal 192. An output filter inductor 202 has a first terminal connected to the positive output terminal and has a second terminal connected to an output voltage node 204. A second output filter capacitor 206 is connected between the output voltage node and the negative output terminal. The two output filter capacitors and the output filter inductor operate in a conventional manner to filter the pulsing DC voltage on the positive output terminal to produce a relatively smooth DC voltage on the output voltage node.
A DC load (LED load) 210 is connected across the second output filter capacitor 206. A first terminal (+) of the LED load is connected to the output voltage node 204. A second terminal (−) of the LED load is coupled to the negative output terminal 192 such that the DC voltage on the output voltage node is applied to the LED load. The DC voltage on the output voltage node is thus referenced to the output (secondary side) ground reference 194. In the illustrated embodiment, the LED load comprises a plurality of light-emitting diodes (LEDs). The LEDs may be connected in series, connected in parallel, or connected in a series-parallel combination.
The above-described components on the secondary side of the isolation transformer 160 operate as the AC-to-DC converter 104 to convert the AC voltage on the secondary winding 170 of the isolation transformer to a DC voltage applied to the LED load 210. In the illustrated embodiment, increasing the switching frequency of the DC-to-AC inverter decreases the current flowing through the LED load, and decreasing the switching frequency increases the current flowing through the LED load.
The second terminal (−) of the LED load 210 can be connected directly to the negative output terminal 192; however, in the illustrated embodiment, the second terminal of the LED load is connected to a first terminal of a current sensing resistor 220 at a sensing node 222. A second terminal of the current sensing resistor is connected to the negative output terminal. Thus, a load current flows through the LED load from the output voltage node 204 and returns to the negative output terminal via the current sensing resistor. A voltage is generated across the voltage sensing resistor proportional to the current flowing through the resistor and thus proportional to the current flowing through the LED load. The resistance of the current sensing resistor is very low (e.g., in a range from approximately 10 milliohms to approximately 50 milliohms) and the resistance tolerance of the current sensing resistor is very tight (e.g., approximately 1 percent). Thus, the voltage across the current sensing resistor measured at the sensing node represents the current flowing through the resistor very accurately. The low resistance causes the voltage drop across the currently sensing resistor to be very small (e.g., approximately 10 millivolts per ampere of current flowing through the LED load). Thus, the voltage drop across the sensing resistor does not reduce the voltage across the LED load significantly.
The voltage across the current sensing resistor 220 at the sensing node 222 is referenced to the output ground reference 194. The sensed voltage is coupled to a high-impedance input of an operational amplifier (OP AMP) 230. The operational amplifier buffers and amplifies the relatively low voltage. The operational amplifier generates an output voltage on an output. The output voltage from the operational amplifier is an analog voltage proportional to the sensed voltage. Thus, the output voltage is proportional to the current through the LED load 210.
The buffered and amplified voltage from the operational amplifier 230 is provided on a first analog input (IN1) of a microcontroller 240. The microcontroller compares the current represented by the analog input voltage with a desired current through the LED load 210. The desired current is represented by a value IREF on a second analog input (IN2) to the microcontroller. The microcontroller generates a secondary feedback signal (FBs) responsive to a calculated differential between the measured current and the desired current. The calculated differential can be considered to be an error value. The character of the feedback signal produced by the microcontroller is determined by parameters of a control (CTL) input of the switch controller 130 in the DC-to-AC inverter 102. As discussed above, the switch controller is responsive to an analog input signal to control the frequency of the signals applied to the control terminals of the two MOSFETs 120, 124. The microcontroller calculates or looks up a value in response to the calculated current differential and generates a value for the secondary feedback signal. The microcontroller may increase the secondary feedback signal to decrease the frequency if the sensed current is lower than the desired current. The microcontroller may decrease the secondary feedback signal to increase the frequency if the current is greater than the desired current. Accordingly, the microcontroller adjusts the secondary feedback signal to maintain the sensed current approximately equal to the desired current. As described below, the feedback signal is communicated to the control input of the switch controller, which is responsive to the feedback signal to vary the frequency to attempt to reduce the current differential to zero. Other feedback techniques may also be used.
In the embodiment of FIG. 1, the secondary feedback signal FBs is not communicated directly to the control input of the switch controller 130. Rather, the secondary feedback signal is communicated from the output of the microcontroller 240 to an input of an isolator circuit 250. The input of the isolator circuit receives the secondary feedback signal relative to the output (secondary side) ground reference 194. The isolator circuit generates a buffered primary feedback signal (FBP) on an output. The buffered primary feedback signal on the output is generated relative to the input (primary side) ground reference 114. Accordingly, the isolator circuit operates to convert the secondary feedback signal (operating with respect to the secondary side ground reference) to the primary feedback signal (operating with respect to the primary side ground reference) without compromising the isolation between the two ground references. The output of the isolator circuit is provided as the input to the control input of the switch controller.
The isolator circuit 250 may be a conventional isolation circuit. For example, a commercially available HCNR200 optocoupler from Avago Technologies may be used. Other optical isolation circuits from other vendors may also be used. For example, isolation may also be provided by using a second isolation transformer to AC-couple the feedback signal between the secondary section and the primary section of the driver circuit 100.
In operation, the switched DC voltage on the common switching node 122 is AC-coupled to the primary winding 162 of the output transformer 160. Accordingly, an AC voltage is produced on the secondary winding 170 of the output transformer. The magnitude of the AC voltage will vary in accordance with the operating frequency of the switch controller 130. The AC output of the secondary winding is rectified by the full-wave bridge rectifier 198 and filtered by output filter inductor 202 and the two output filter capacitors 200, 206 to produce the DC voltage to drive the LED load 210. The current through the LED load is sensed as a voltage across the sensing resistor 220. The sensed voltage is buffered and amplified by the operational amplifier 230. The microcontroller 240 compares the current represented by the sensed voltage with a desired current (a reference current) and generates the secondary feedback signal (FBS) responsive to a calculated difference between the two currents. The isolator circuit 250 converts the secondary feedback signal (referenced to the output ground reference 194) to the primary feedback signal FBP (referenced to the input ground reference 114). The primary feedback signal is applied to the control (CTL) input of the switch controller, which is responsive to the feedback signal to adjust the operating frequency of the signals applied to the control terminals of the two MOSFETs. Adjusting the operating frequency adjusts the power transferred to the secondary side of the driver circuit 100 and thus adjusts the magnitude of the current flowing through the LED load. The magnitude of the current is adjusted to reduce the calculated current difference. For example, if the sensed current is too low, the frequency is adjusted to transfer more energy into the tank circuit to increase the DC voltage and thereby increase the current through the LED load. If the sensed current is too high, the frequency is adjusted to transfer less energy into the tank circuit to decrease the DC voltage and thereby decrease the current through the LED load.